Filter circuit, wireless transmitter, and wireless receiver

Abstract

A filter circuit is constructed of a passive mixer and a frequency response device. The passive mixer generates a frequency-converted signal by mixing an input signal transmitted through a transmission line and a local oscillation signal, and outputs the frequency-converted signal to the frequency response device, thus shifting a frequency characteristic of the frequency response device to high frequency by a local oscillation frequency of the local oscillation signal and applying the frequency characteristic shifted to high frequency on the input signal.

Claims

1 . A filter circuit comprising: a frequency response device having a frequency characteristic; and a passive mixer that mixes an input signal and a local oscillation signal, and outputs a mixed signal to the frequency response device, thereby shifting the frequency characteristic of the frequency response device to high frequency by a local oscillation frequency of the local oscillation signal and applying the frequency characteristic shifted to high frequency on the input signal. 2 . The filter circuit according to claim 1 , wherein an input terminal of the passive mixer is connected to a transmission line through which the input signal is transmitted, and an output terminal of the passive mixer is connected to the frequency response device. 3 . The filter circuit according to claim 2 , wherein the passive mixer includes a field effect transistor, a drain of the field effect transistor is connected to the transmission line, a source of the field effect transistor is connected to the frequency response device, and the local oscillation signal is input to a gate of the field effect transistor. 4 . The filter circuit according to claim 2 , wherein the input signal is a differential signal, and the passive mixer mixes the input signal and local oscillation signals having different phases by 180 degrees. 5 . The filter circuit according to claim 4 , wherein the passive mixer is a differential passive mixer, and frequency-converted signals output from the differential passive mixer are input to terminals of the frequency response device, respectively. 6 . The filter circuit according to claim 5 , wherein the passive mixer includes a first field effect transistor and a second field effect transistor, a drain of the first field effect transistor is connected to a first transmission line through which a first signal of the differential signal is transmitted, a source of the first field effect transistor is connected to a first terminal of the frequency response device, a drain of the second field effect transistor is connected to a second transmission line through which a second signal of the differential signal is transmitted, a source of the second field effect transistor is connected to a second terminal of the frequency response device, and a phase of a local oscillation signal input to a gate of the first field effect transistor is different from a phase of a local oscillation signal input to a gate of the second field effect transistor by 180 degrees. 7 . The filter circuit according to claim 5 , wherein the passive mixer includes a first field effect transistor, a second field effect transistor, a third field effect transistor, and a fourth field effect transistor, a drain of the first field effect transistor is connected to a first transmission line through which a first signal of the differential signal is transmitted, a source of the first field effect transistor is connected to a first terminal of the frequency response device, a drain of the second field effect transistor is connected to the drain of the first field effect transistor, a source of the second field effect transistor is connected to a second terminal of the frequency response device, a drain of the third frequency response device is connected to a second transmission line through which a second signal of the differential signal is transmitted, a source of the third field effect transistor is connected to the source of the first field effect transistor, a drain of the fourth field effect transistor is connected to the drain of the third field effect transistor, a source of the fourth field effect transistor is connected to the source of the second field effect transistor, and a phase of a local oscillation signal input to gates of the first field effect transistor and the fourth field effect transistor is different from a phase of a local oscillation signal input to gates of the second field effect transistor and the third field effect transistor by 180 degrees. 8 . The filter circuit according to claim 4 , wherein the passive mixer is a differential passive mixer, and the frequency response device includes a first frequency response device to which a first frequency-converted signal output from the differential passive mixer is input, and a second frequency response device to which a second frequency-converted signal output from the differential passive mixer is input. 9 . The filter circuit according to claim 1 , wherein the passive mixer includes a first passive mixer for an inphase signal and a second passive mixer for an quadrature phase signal, the first passive mixer and the second passive mixer mix the input signal and local oscillation signals having different phases by 90 degrees, respectively. 10 . The filter circuit according to claim 9 , wherein the input signal is a differential signal, and each of the first passive mixer and the second passive mixer is a single-phase passive mixer. 11 . The filter circuit according to claim 10 , wherein a first signal of the differential signal is input to the first passive mixer, a second signal of the differential signal is input to the second passive mixer, a frequency-converted signal output from the first passive mixer is input to a first terminal of the frequency response device, and a frequency-converted signal output from the second passive mixer is input to a second terminal of the frequency response device. 12 . The filter circuit according to claim 11 , wherein the passive mixer includes a first field effect transistor, a second field effect transistor, a third field effect transistor, and a fourth field effect transistor, a drain of the first field effect transistor is connected to a first transmission line through which the first signal of the differential signal is transmitted, a source of the first field effect transistor is connected to the first terminal of the frequency response device, a drain of the second field effect transistor is connected to the drain of the first field effect transistor, a source of the second field effect transistor is connected to the second terminal of the frequency response device, a drain of the third frequency response device is connected to a second transmission line through which the second signal of the differential signal is transmitted, a source of the third field effect transistor is connected to the source of the first field effect transistor, a drain of the fourth field effect transistor is connected to the drain of the third field effect transistor, a source of the fourth field effect transistor is connected to the source of the second field effect transistor, and phases of local oscillation signals input to gates of the first field effect transistor, the second field effect transistor, the third field effect transistor, and the fourth field effect transistor are sequentially different from each other by 90 degrees. 13 . The filter circuit according to claim 10 , wherein a first signal of the differential signal is input to the first passive mixer, a second signal of the differential signal is input to the second passive mixer, and the frequency response device includes a first frequency response device to which a frequency-converted signal output from the first passive mixer is input, and a second frequency response device to which a frequency-converted signal output from the second passive mixer is input. 14 . The filter circuit according to claim 9 , wherein the input signal is a differential signal, and each of the first passive mixer and the second passive mixer is a differential passive mixer. 15 . The filter circuit according to claim 14 , wherein the passive mixer is a differential passive mixer, and the frequency response device includes a first frequency response device to which a pair of first frequency-converted signals output from the first passive mixer is input, and a second frequency response device to which a pair of second frequency-converted signals output from the second passive mixer is input. 16 . The filter circuit according to claim 14 , wherein the frequency response device includes a first frequency response device to which a first frequency-converted signal output from the first passive mixer is input, a second frequency response device to which a second frequency-converted signal output from the first passive mixer is input, a third frequency response device to which a first frequency-converted signal output from the second passive mixer is input, and a fourth frequency response device to which a second frequency-converted signal output from the second passive mixer is input. 17 . A radio transmitter comprising: an up-converter that converts a baseband signal or an intermediate-frequency signal into a radio-frequency signal; a frequency response device having a frequency characteristic; a passive mixer that mixes the radio-frequency signal and a local oscillation signal, and outputs a mixed signal to the frequency response device, thereby shifting the frequency characteristic of the frequency response device to high frequency by a local oscillation frequency of the local oscillation signal and applying the frequency characteristic shifted to high frequency on the radio-frequency signal; and a transmission antenna that transmits the radio-frequency signal on which the frequency characteristic of the frequency response device shifted to high frequency is applied to a space. 18 . The radio transmitter according to claim 17 , wherein an input terminal of the passive mixer is connected to a transmission line through which the radio-frequency signal is transmitted, and an output terminal of the passive mixer is connected to the frequency response device. 19 . A radio receiver comprising: a reception antenna that receives a radio-frequency signal propagating in a space; a frequency response device having a frequency characteristic; a passive mixer that mixes the radio-frequency signal and a local oscillation signal, and outputs a mixed signal to the frequency response device, thereby shifting the frequency characteristic of the frequency response device to high frequency by a local oscillation frequency of the local oscillation signal and applying the frequency characteristic shifted to high frequency on the radio-frequency signal; and a down-converter that converts the radio-frequency signal on which the frequency characteristic of the frequency response device shifted to high frequency is applied into a baseband signal or an intermediate-frequency signal. 20 . The radio receiver according to claim 19 , wherein an input terminal of the passive mixer is connected to a transmission line through which the radio-frequency signal is transmitted, and an output terminal of the passive mixer is connected to the frequency response device.
CROSS-REFERENCE TO RELATED APPLICATIONS [0001] This application is based upon and claims the benefit of priority from the prior Japanese Patent Application No. 2008-131837, filed on May 20, 2008; the entire contents of which are incorporated herein by reference. BACKGROUND OF THE INVENTION [0002] 1. Field of the Invention [0003] The present invention relates to a filter circuit, a wireless transmitter, and a wireless receiver, and more particularly, to a filter circuit including a passive mixer. [0004] 2. Description of the Related Art [0005] Generally, a wireless communication apparatus is equipped with a filter circuit for selecting a signal of a specific frequency (see, for example, Japanese Patent Application Laid-open No. 2005-130381). Such a filter circuit is built by connecting a parallel circuit constructed of an inductor and a capacitor to a transmission line. The resonant frequency f of the filter circuit is given by f=1/(2π(LC) 1/2 ), where L [Henry] is the inductance of the inductor and C [Farad] is the capacitance of the capacitor. [0006] If a frequency of a signal passing the transmission line is close to the resonant frequency, the impedance of the filter circuit becomes close to infinity, and as the frequency of the signal passing the transmission line goes apart from the resonant frequency, the impedance of the filter circuit decreases. Therefore, the filter circuit does not attenuate a signal having a frequency close to the resonant frequency, while attenuating a signal having a frequency apart from the resonant frequency, which makes it possible to select the signal having the frequency close to the resonant frequency. [0007] When a variable inductor or a variable capacitor is used in the filter circuit, the resonant frequency can be changed by changing the inductance L or the capacitance, so that the signal attenuation band can be changed. [0008] However, in a conventional filter circuit, because it becomes susceptible to an influence of the parasitic capacitance or the parasitic inductance as the resonant frequency (i.e., the filter center frequency) increases, it is hard to precisely match the resonant frequency to a target value by an electronic control, and it is hard to obtain a steep attenuation characteristic or a large attenuation amount. [0009] Particularly when making the attenuation characteristic steep or increasing the attenuation amount, it is indispensible to precisely match the resonant frequency. However, with a frequency near 2 gigahertz, for example, it is hard to precisely match the resonant frequency on the order of 100 kilohertz. BRIEF SUMMARY OF THE INVENTION [0010] A filter circuit according to an embodiment of the present invention comprises: a frequency response device having a frequency characteristic; and a passive mixer that mixes an input signal and a local oscillation signal, and outputs a mixed signal to the frequency response device, thereby shifting the frequency characteristic of the frequency response device to high frequency by a local oscillation frequency of the local oscillation signal and applying the frequency characteristic shifted to high frequency on the input signal. [0011] A radio transmitter according to an embodiment of the present invention comprises: an up-converter that converts a baseband signal or an intermediate-frequency signal into a radio-frequency signal; a frequency response device having a frequency characteristic; a passive mixer that mixes the radio-frequency signal and a local oscillation signal, and outputs a mixed signal to the frequency response device, thereby shifting the frequency characteristic of the frequency response device to high frequency by a local oscillation frequency of the local oscillation signal and applying the frequency characteristic shifted to high frequency on the radio-frequency signal; and a transmission antenna that transmits the radio-frequency signal on which the frequency characteristic of the frequency response device shifted to high frequency is applied to a space. [0012] A radio receiver according to an embodiment of the present invention comprises: a reception antenna that receives a radio-frequency signal propagating in a space; a frequency response device having a frequency characteristic; a passive mixer that mixes the radio-frequency signal and a local oscillation signal, and outputs a mixed signal to the frequency response device, thereby shifting the frequency characteristic of the frequency response device to high frequency by a local oscillation frequency of the local oscillation signal and applying the frequency characteristic shifted to high frequency on the radio-frequency signal; and a down-converter that converts the radio-frequency signal on which the frequency characteristic of the frequency response device shifted to high frequency is applied into a baseband signal or an intermediate-frequency signal. BRIEF DESCRIPTION OF THE DRAWINGS [0013] FIG. 1 is a block diagram of a filter circuit according to a first embodiment of the present invention; [0014] FIG. 2 is a circuit diagram illustrating an example of a frequency response device composed of a capacitor for the filter circuit according to the first embodiment; [0015] FIG. 3 is a circuit diagram illustrating another example of the frequency response device composed of an inductor for the filter circuit according to the first embodiment; [0016] FIG. 4 is a circuit diagram illustrating still another example of the frequency response device composed of a capacitor and an inductor connected in series for the filter circuit according to the first embodiment; [0017] FIG. 5 is a circuit diagram illustrating still another example of the frequency response device composed of a capacitor and an inductor connected in parallel for the filter circuit according to the first embodiment; [0018] FIG. 6 is a block diagram of a filter circuit according to a second embodiment of the present invention; [0019] FIG. 7 is a block diagram of a filter circuit according to a third embodiment of the present invention; [0020] FIG. 8 is a block diagram of a filter circuit according to a fourth embodiment of the present invention; [0021] FIG. 9 is a block diagram of a filter circuit according to a fifth embodiment of the present invention; [0022] FIG. 10 is a block diagram of a filter circuit according to a sixth embodiment of the present invention; [0023] FIG. 11 is a block diagram of a filter circuit according to a seventh embodiment of the present invention; [0024] FIG. 12 is a block diagram of a filter circuit according to an eighth embodiment of the present invention; [0025] FIG. 13 is a block diagram of a filter circuit according to a ninth embodiment of the present invention; [0026] FIG. 14 is a block diagram of a filter circuit according to a tenth embodiment of the present invention; [0027] FIG. 15 is a block diagram of a filter circuit according to an eleventh embodiment of the present invention; [0028] FIG. 16 is a block diagram of a filter circuit according to a twelfth embodiment of the present invention; [0029] FIG. 17 is a block diagram of a filter circuit according to a thirteenth embodiment of the present invention; [0030] FIG. 18 is a block diagram of a passive mixer applied to a filter circuit according to a fourteenth embodiment of the present invention; [0031] FIG. 19 is a block diagram of a passive mixer applied to a filter circuit according to a fifteenth embodiment of the present invention; [0032] FIG. 20 is a block diagram of a passive mixer applied to a filter circuit according to a sixteenth embodiment of the present invention; [0033] FIG. 21 is a block diagram of a passive mixer applied to a filter circuit according to a seventeenth embodiment of the present invention; [0034] FIG. 22 is a block diagram of a passive mixer applied to a filter circuit according to an eighteenth embodiment of the present invention; [0035] FIG. 23 is a block diagram of a passive mixer applied to a filter circuit according to a nineteenth embodiment of the present invention; [0036] FIG. 24 is a block diagram of a passive mixer applied to a filter circuit according to a twentieth embodiment of the present invention; [0037] FIG. 25 is a block diagram of a passive mixer applied to a filter circuit according to a twenty-first embodiment of the present invention; [0038] FIG. 26 is a block diagram of a passive mixer applied to a filter circuit according to a twenty-second embodiment of the present invention; [0039] FIG. 27 is a block diagram of a wireless transmitter according to a twenty-third embodiment of the present invention to which the filter circuit according to the present invention is applied; and [0040] FIG. 28 is a block diagram of a wireless receiver according to a twenty-fourth embodiment of the present invention to which the filter circuit according to the present invention is applied. DETAILED DESCRIPTION OF THE INVENTION [0041] Exemplary embodiments of a filter circuit, a wireless transmitter, and a wireless receiver according to the present invention will be explained in detail below with reference to the accompanying drawings. The present invention is not limited to the following embodiments. [0042] FIG. 1 is a block diagram of a filter circuit according to a first embodiment of the present invention. [0043] As shown in FIG. 1 , the filter circuit according to the first embodiment includes a passive mixer 2 and a frequency response device 3 . The input terminal of the passive mixer 2 is connected to a transmission line 1 , and the output terminal of the passive mixer 2 is grounded via the frequency response device 3 . The frequency response device 3 can be constructed of a device having some sort of frequency response F. For example, the frequency response device 3 can be constructed of a capacitor C 1 shown in FIG. 2 , an inductor L 1 shown in FIG. 3 , a serial circuit constructed of a capacitor C 2 and an inductor L 2 shown in FIG. 4 , and a parallel circuit constructed of a capacitor C 3 and an inductor L 3 shown in FIG. 5 . [0044] The passive mixer 2 can generate a frequency-converted signal B by mixing an input signal T that is transmitted through the transmission line 1 and a local oscillation signal LO. In doing so, the passive mixer 2 can apply the frequency characteristic F of the frequency response device 3 , which is shifted to high frequency by a local oscillation frequency f LO of the local oscillation signal LO, on the input signal T by outputting the frequency-converted signal B to the frequency response device 3 . The passive mixer 2 is a mixer that does not have an amplification function, and can be constructed of, for example, a mechanical switch or the like, such as a relay switch, or a transistor or the like that performs an on/off switching operation. In the case of using a transistor for the passive mixer 2 , the transistor can be used as a variable resistor, and therefore, the input signal T can be applied to the source or the drain without being applied to the gate. [0045] In other words, by mixing the input signal T and the local oscillation signal LO at the passive mixer 2 , the frequency-converted signal B having frequencies of f BB1 =|f RF −f LO | and f BB2 =|f RF +f LO | is generated, where f RF is the frequency of the input signal T that is transmitted through the transmission line 1 , and the frequency-converted signal B is output to the frequency response device 3 . When the frequency-converted signal B is input to the frequency response device 3 , the frequency characteristic F of the frequency response device 3 is applied on the frequency-converted signal B. [0046] Considering the frequency f BB1 , when the frequency characteristic F of the frequency response device 3 is applied on the frequency-converted signal B, it means that a frequency characteristic F′, which is the frequency characteristic F shifted to high frequency by the local oscillation frequency f LO , is applied on the input signal T. [0047] Therefore, for the input signal T, the passive mixer 2 and the frequency response device 3 can operate as a filter circuit that has the frequency characteristic F′ which is the frequency characteristic F shifted to high frequency by the local oscillation frequency f LO . Furthermore, the center frequency of this filter circuit can be easily changed by adjusting the local oscillation frequency f LO of the local oscillation signal LO. [0048] The local oscillation frequency f LO of the local oscillation signal LO can be precisely adjusted on the order of 100 kilohertz even at a high frequency near 2 gigahertz by using a phase-locked loop (PLL) circuit, a delay-locked loop (DLL) circuit, or the like. As a result, even when the frequency of the input signal T is high, the filter center frequency for the input signal T can be adjusted using the frequency response device 3 that has a low filter frequency, which makes it possible to precisely adjust the center frequency of the filter circuit, and to make the attenuation characteristic steep and increase the attenuation amount for the input signal T. [0049] FIG. 6 is a block diagram of a filter circuit according to a second embodiment of the present invention. [0050] As shown in FIG. 6 , the filter circuit according to the second embodiment includes a passive mixer 2 ′ and frequency response devices 3 a and 3 b. The input terminal of the passive mixer 2 ′ is connected to a transmission line 1 , and the output terminal of the passive mixer 2 ′ is grounded via the frequency response devices 3 a and 3 b. Each of the frequency response devices 3 a and 3 b can be constructed of a device having some sort of frequency response F. For example, each of the frequency response devices 3 a and 3 b can be constructed of the capacitor C 1 shown in FIG. 2 , the inductor L 1 shown in FIG. 3 , the serial circuit constructed of the capacitor C 2 and the inductor L 2 shown in FIG. 4 , and the parallel circuit constructed of the capacitor C 3 and the inductor L 3 shown in FIG. 5 . [0051] The passive mixer 2 ′ can generate frequency-converted signals B 1 + and B 1 − by mixing an input signal T that is transmitted through the transmission line 1 and local oscillation signals LO + and LO − . As for the local oscillation signals LO + and LO − , signals having the same frequency with opposite phases can be used. The passive mixer 2 ′ can shift the frequency characteristic F of the frequency response devices 3 a and 3 b to high frequency by a local oscillation frequency f LO of the local oscillation signals LO + and LO − by outputting the frequency-converted signals B 1 + and B 1 − to the frequency response devices 3 a and 3 b, respectively, and thus applying the frequency characteristic F that is shifted to high frequency on the input signal T. [0052] In other words, by applying the frequency characteristic F of the frequency response devices 3 a and 3 b on the frequency-converted signals B 1 + and B 1 − , respectively, it is possible to apply a frequency characteristic F′ which is the frequency characteristic F shifted to high frequency by the local oscillation frequency f LO , on the input signal T. [0053] Therefore, for the input signal T, the passive mixer 2 ′ and the frequency response devices 3 a and 3 b can operate as a filter circuit that has the frequency characteristic F′ which is the frequency characteristic F shifted to high frequency by the local oscillation frequency f LO . As a result, even when the frequency of the input signal T is high, the center frequency of the filter circuit can be precisely matched to a target value, and at the same time, it is possible to make the attenuation characteristic steep and increase the attenuation amount for the input signal T. [0054] FIG. 7 is a block diagram of a filter circuit according to a third embodiment of the present invention. [0055] As shown in FIG. 7 , the filter circuit according to the third embodiment includes a passive mixer 12 and a frequency response device 3 . As for the passive mixer 12 , a differential passive mixer can be used, in which a pair of input terminals for inputting differentials signal, a pair of control terminals for inputting local oscillation signals LO + and LO − , and a pair of output terminals for outputting frequency-converted signals can be provided. [0056] The input terminals of the passive mixer 12 are connected to transmission lines 1 a and 1 b, respectively, and the output terminals of the passive mixer 12 are connected to each other via the frequency response device 3 . [0057] The passive mixer 12 can generate frequency-converted signals B 2 + and B 2 − by mixing input signals T + and T − that are transmitted through the transmission lines 1 a and 1 b and the local oscillation signals LO + and LO − , respectively. The transmission lines 1 a and 1 b can build a differential transmission line, and as for the input signals T + and T − , a differential signal can be used. The passive mixer 12 can shift a frequency characteristic F of the frequency response device 3 to high frequency by a local oscillation frequency f LO of the local oscillation signals LO + and LO − by outputting the frequency-converted signals B 2 + and B 2 − to both terminals of the frequency response device 3 , respectively, and apply the frequency characteristic F that is shifted to high frequency on the input signals T + and T − . [0058] In other words, by applying the frequency characteristic F of the frequency response device 3 on the frequency-converted signals B 2 + and B 2 − , it is possible to apply a frequency characteristic F′ which is the frequency characteristic F shifted to high frequency by the local oscillation frequency f LO , on the input signals T + and T − . [0059] Therefore, for the input signals T + and T − , the passive mixer 12 and the frequency response device 3 can operate as a filter circuit that has the frequency characteristic F′ which is the frequency characteristic F shifted to high frequency by the local oscillation frequency f LO . As a result, even when a high-frequency differential signal is input as the input signals T + and T − , the center frequency of the filter circuit can be precisely matched to a target value, and at the same time, it is possible to make the attenuation characteristic steep and increase the attenuation amount for the input signals T + and T − , and to increase the signal transmission speed while decreasing the common-mode noise. [0060] FIG. 8 is a block diagram of a filter circuit according to a fourth embodiment of the present invention with a configuration that the frequency response device shown in FIG. 3 is applied to the filter circuit shown in FIG. 7 . [0061] As shown in FIG. 8 , the capacitor C 1 is used as the frequency response device 3 of the filter circuit shown in FIG. 7 . A frequency characteristic F 1 of the capacitor C 1 is that the impedance Z increase as the frequency f decreases and the impedance Z becomes infinity at the frequency zero (direct current). The impedance Z decreases as the frequency f increase. Therefore, when the frequency f BB1 of the frequency-converted signals B 2 + and B 2 − is low, less current flows in the capacitor C 1 , and on the other hand, when the frequency f BB1 of the frequency-converted signals B 2 + and B 2 − is high, more current flows in the capacitor C 1 . [0062] When the input terminals of the passive mixer 12 are connected to the transmission lines 1 a and 1 b, respectively, the impedance characteristic of the output terminals of the passive mixer 12 can be observed from the transmission lines 1 a and 1 b side, too. In the case in which more current flows at the output terminals of the passive mixer 12 (i.e., when the frequency f RF of the input signals T + and T − is apart from the local oscillation frequency f LO ), more current flows at the transmission lines 1 a and 1 b side, too. On the other hand, in the case in which less current flows at the output terminals of the passive mixer 12 (i.e., when the frequency f RF of the input signals T + and T − is close to the local oscillation frequency f LO ), less current flows at the transmission lines 1 a and 1 b side, too. [0063] From the above aspect, when the frequency f RF of the input signals T + and T − is apart from the local oscillation frequency f LO , the current is easily flown to the passive mixer 12 from the transmission lines 1 a and 1 b, and when the frequency f RF of the input signals T + and T − is close to the local oscillation frequency f LO , the current is hardly flown to the passive mixer 12 from the transmission lines 1 a and 1 b. As a result, the frequency characteristic F 1 of the capacitor C 1 , which is symmetrically folded at the local oscillation frequency f LO as the center, is up-converted to the transmission lines 1 a and 1 b side by the local oscillation frequency f LO , and it is possible to make a band-pass characteristic F 1 ′ with the center at the local oscillation frequency f LO appear on the transmission lines 1 a and 1 b side. [0064] Considering the frequency f BB2 of the frequency-converted signals B 2 + and B 2 − , both when the frequency f RF of the input signals T + and T − is apart from the local oscillation frequency f LO and when the frequency f RF of the input signals T + and T − is close to the local oscillation frequency f LO , the frequency f BB2 of the frequency-converted signals B 2 + and B 2 − becomes high, and the impedance Z of the capacitor C 1 is decreased. [0065] For this reason, for the frequency f BB2 , a large change does not occur in the impedance Z on the frequency characteristic F 1 of the capacitor C 1 , and for the frequency f BB1 , a large change occurs in the impedance Z on the frequency characteristic F 1 of the capacitor C 1 . Therefore, when the total impedance characteristic of the output terminals of the passive mixer 12 is observed from the transmission lines 1 a and 1 b side, the change of the impedance Z can be ignored on the frequency characteristic F 1 in the frequency f BB2 , and only the change of the impedance Z on the frequency characteristic F 1 in the frequency f BB1 can be dominantly observed as the total change of the impedance Z. [0066] As a result, by using the capacitor C 1 as the frequency response device 3 of the filter circuit shown in FIG. 7 , it is possible to obtain the band-pass characteristic F 1 ′ with the local oscillation frequency f LO as the center with respect to transmission lines 1 a and 1 b. Because the local oscillation frequency f LO can be freely varied, the center frequency of the pass band can be also changed freely and precisely. Furthermore, because the Q factor of the capacitor C 1 can be easily increased, a band-pass filter having an extremely narrow bandwidth can be constructed. In addition, because the passive mixer 12 can be easily formed by a semiconductor manufacturing process, it can be constructed on the same integrated circuit (IC) chip with the other circuits, which leads to a cost down without using an external filter device. Moreover, since it does not consume the electric power because it is a passive mixer, the filter function can be added without increasing the power consumption. As a result, by using the passive mixer 12 and the capacitor C 1 , it is possible to realize a high-performance tunable band-pass filter with ease. [0067] FIG. 9 is a block diagram of a filter circuit according to a fifth embodiment of the present invention with a configuration that the frequency response device shown in FIG. 4 is applied to the filter circuit shown in FIG. 7 . [0068] As shown in FIG. 9 , the inductor L 1 is used as the frequency response device 3 of the filter circuit shown in FIG. 7 . The frequency characteristic F 2 of the inductor L 1 is that the impedance Z decrease as the frequency f decreases and the impedance Z becomes zero at the frequency zero (direct current). The impedance Z increase as the frequency f increases. Therefore, when the frequency f BB1 of the frequency-converted signals B 2 + and B 2 − is low, more current flows in the inductor L 1 , and on the other hand, when the frequency f BB1 of the frequency-converted signals B 2 + and B 2 − is high, less current flows in the inductor L 1 . [0069] When the input terminals of the passive mixer 12 are connected to the transmission lines 1 a and 1 b, respectively, the impedance characteristic of the output terminals of the passive mixer 12 can be observed from the transmission lines 1 a and 1 b side, too. In the case in which more current flows at the output terminals of the passive mixer 12 (i.e., when the frequency f RF of the input signals T + and T − is close to the local oscillation frequency f LO ), more current flows at the transmission lines 1 a and 1 b side, too. On the other hand, in the case in which less current flows at the output terminals of the passive mixer 12 (i.e., when the frequency f RF of the input signals T + and T is apart from the local oscillation frequency f LO ), less current flows at the transmission lines 1 a and 1 b side, too. [0070] From the above aspect, when the frequency f RF of the input signals T + and T − is close to the local oscillation frequency f LO , the current is easily flown to the passive mixer 12 from the transmission lines 1 a and 1 b, and when the frequency f RF of the input signals T + and T − is apart from the local oscillation frequency f LO , the current is hardly flown to the passive mixer 12 from the transmission lines 1 a and 1 b. As a result, the frequency characteristic F 2 of the inductor L 1 , which is symmetrically folded at the local oscillation frequency f LO as the center, is up-converted to the transmission lines 1 a and 1 b side by the local oscillation frequency f LO , and it is possible to make a band-stop characteristic F 2 ′ with the center at the local oscillation frequency f LO appear on the transmission lines 1 a and 1 b side. [0071] Considering the frequency f BB2 of the frequency-converted signals B 2 + and B 2 − , both when the frequency f RF of the input signals T + and T is apart from the local oscillation frequency f LO and when the frequency f RF of the input signals T + and T − is close to the local oscillation frequency f LO , the frequency f BB2 of the frequency-converted signals B 2 + and B 2 − becomes high, and the impedance Z of the inductor L 1 is increased. [0072] For this reason, for the frequency f BB2 , a large change does not occur in the impedance Z on the frequency characteristic F 2 of the inductor L 1 , and for the frequency f BB1 , a large change occurs in the impedance Z on the frequency characteristic F 2 of the inductor L 1 . Therefore, when the total impedance characteristic of the output terminals of the passive mixer 12 is observed from the transmission lines 1 a and 1 b side, the change of the impedance Z can be ignored on the frequency characteristic F 2 in the frequency f BB2 , and only the change of the impedance Z on the frequency characteristic F 2 in the frequency f BB1 can be dominantly observed as the total change of the impedance Z. [0073] As a result, by using the inductor L 1 as the frequency response device 3 of the filter circuit shown in FIG. 7 , it is possible to obtain the band-stop characteristic F 2 ′ with the local oscillation frequency f LO as the center with respect to transmission lines 1 a and 1 b. Because the local oscillation frequency f LO can be freely varied, the center frequency of the pass band can be also changed freely and precisely. Furthermore, because the Q factor of the inductor L 1 can be easily increased, a band-stop filter having an extremely narrow bandwidth can be constructed. In addition, because the passive mixer 12 can be easily formed by a semiconductor manufacturing process, it can be constructed on the same integrated circuit (IC) chip with the other circuits, which leads to a cost down without using an external filter device. Moreover, since it does not consume the electric power because it is a passive mixer, the filter function can be added without increasing the power consumption. As a result, by using the passive mixer 12 and the inductor L 1 , it is possible to realize a high-performance tunable band-stop filter with ease. [0074] FIG. 10 is a block diagram of a filter circuit according to a sixth embodiment of the present invention with a configuration that the frequency response device shown in FIG. 5 is applied to the filter circuit shown in FIG. 7 . [0075] As shown in FIG. 10 , the serial circuit of the inductor L 2 and the capacitor C 2 is used as the frequency response device 3 of the filter circuit shown in FIG. 7 . The resonant frequency of the serial circuit of the inductor L 2 and the capacitor C 2 is f R , and it is assumed that the resonant frequency f R is lower enough than the local oscillation frequency f LO . A frequency characteristic F 3 of the serial circuit of the inductor L 2 and the capacitor C 2 is that the impedance Z becomes low near the resonant frequency f R , and the impedance Z increase as the frequency is apart from resonant frequency f R . Therefore, when the frequency f BB1 of the frequency-converted signals B 2 + and B 2 − is near the resonant frequency f R , more current flows in the serial circuit of the inductor L 2 and the capacitor C 2 , and on the other hand, when the frequency f BB1 of the frequency-converted signals B 2 + and B 2 − is apart from the resonant frequency f R , less current flows in the serial circuit of the inductor L 2 and the capacitor C 2 . [0076] When the input terminals of the passive mixer 12 are connected to the transmission lines 1 a and 1 b, respectively, the impedance characteristic of the output terminals of the passive mixer 12 can be observed from the transmission lines 1 a and 1 b side, too. In the case in which more current flows at the output terminals of the passive mixer 12 (i.e., when the frequency f RF of the input signals T + and T − is close to f LO ±f R ), more current flows at the transmission lines 1 a and 1 b side, too. On the other hand, in the case in which less current flows at the output terminals of the passive mixer 12 (i.e., when the frequency f RF of the input signals T + and T − is apart from f LO ±f R ), less current flows at the transmission lines 1 a and 1 b side, too. [0077] From the above aspect, when the frequency f RF of the input signals T + and T − is close to f LO ±f R , the current is easily flown to the passive mixer 12 from the transmission lines 1 a and 1 b, and when the frequency f RF of the input signals T + and T − is apart from f LO ±f R , the current is hardly flown to the passive mixer 12 from the transmission lines 1 a and 1 b. As a result, the frequency characteristic F 3 of the serial circuit of the inductor L 2 and the capacitor C 2 , which is symmetrically folded at the local oscillation frequency f LO as the center, is up-converted to the transmission lines 1 a and 1 b side by the local oscillation frequency f LO , and it is possible to make a two-band band-stop characteristic F 3 ′ with the center at the local oscillation frequency f LO appear on the transmission lines 1 a and 1 b side. [0078] Considering the frequency f BB2 of the frequency-converted signals B 2 + and B 2 − , both when the frequency f RF of the input signals T + and T − is apart from f LO ±f R and when the frequency f RF of the input signals T + and T − is close to f LO ±f R , the frequency f BB2 of the frequency-converted signals B 2 + and B 2 − becomes high, and the impedance Z of the serial circuit of the inductor L 2 and the capacitor C 2 is increased. [0079] For this reason, for the frequency f BB2 , a large change does not occur in the impedance Z on the frequency characteristic F 3 of the serial circuit of the inductor L 2 and the capacitor C 2 , and for the frequency f BB1 , a large change occurs in the impedance Z on the frequency characteristic F 3 of the serial circuit of the inductor L 2 and the capacitor C 2 . Therefore, when the total impedance characteristic of the output terminals of the passive mixer 12 is observed from the transmission lines 1 a and 1 b side, the change of the impedance Z can be ignored on the frequency characteristic F 3 in the frequency f BB2 , and only the change of the impedance Z on the frequency characteristic F 3 in the frequency f BB1 can be dominantly observed as the total change of the impedance Z. [0080] As a result, by using the serial circuit of the inductor L 2 and the capacitor C 2 as the frequency response device 3 of the filter circuit shown in FIG. 7 , it is possible to obtain the band-stop characteristic F 3 ′ with f LO ±f R as the center with respect to transmission lines 1 a and 1 b. Because the local oscillation frequency f LO can be freely varied, the center frequency of the pass band can be also changed freely and precisely. Furthermore, because the Q factors of the inductor L 2 and the capacitor C 2 can be easily increased, a band-stop filter having an extremely narrow bandwidth can be constructed. In addition, because the passive mixer 12 can be easily formed by a semiconductor manufacturing process, it can be constructed on the same integrated circuit (IC) chip with the other circuits, which leads to a cost down without using an external filter device. Moreover, since it does not consume the electric power because it is a passive mixer, the filter function can be added without increasing the power consumption. As a result, by using the passive mixer 12 and the serial circuit of the inductor L 2 and the capacitor C 2 , it is possible to realize a high-performance tunable band-stop filter with ease. [0081] FIG. 11 is a block diagram of a filter circuit according to a seventh embodiment of the present invention with a configuration that the frequency response device shown in FIG. 6 is applied to the filter circuit shown in FIG. 7 . [0082] As shown in FIG. 11 , the parallel circuit of the inductor L 3 and the capacitor C 3 is used as the frequency response device 3 of the filter circuit shown in FIG. 7 . The resonant frequency of the parallel circuit of the inductor L 3 and the capacitor C 3 is f R , and it is assumed that the resonant frequency f R is lower enough than the local oscillation frequency f LO . A frequency characteristic F 4 of the parallel circuit of the inductor L 3 and the capacitor C 3 is that the impedance Z becomes high near the resonant frequency f R , and the impedance Z decreases as the frequency is apart from resonant frequency f R . Therefore, when the frequency f BB1 of the frequency-converted signals B 2 + and B 2 − is apart from the resonant frequency f R , more current flows in the parallel circuit of the inductor L 3 and the capacitor C 3 , and on the other hand, when the frequency f BB1 of the frequency-converted signals B 2 + and B 2 − is close to the resonant frequency f R , less current flows in the parallel circuit of the inductor L 3 and the capacitor C 3 . [0083] When the input terminals of the passive mixer 12 are connected to the transmission lines 1 a and 1 b, respectively, the impedance characteristic of the output terminals of the passive mixer 12 can be observed from the transmission lines 1 a and 1 b side, too. In the case in which more current flows at the output terminals of the passive mixer 12 (i.e., when the frequency f RF of the input signals T + and T − is apart from f LO ±f R ), more current flows at the transmission lines 1 a and 1 b side, too. On the other hand, in the case in which less current flows at the output terminals of the passive mixer 12 (i.e., when the frequency f RF of the input signals T + and T − is close to f LO ±f R ), less current flows at the transmission lines 1 a and 1 b side, too. [0084] From the above aspect, when the frequency f RF of the input signals T + and T − is apart from f LO ±f R , the current is easily flown to the passive mixer 12 from the transmission lines 1 a and 1 b, and when the frequency f RF of the input signals T + and T − is close to f LO ±f R , the current is hardly flown to the passive mixer 12 from the transmission lines 1 a and 1 b. As a result, the frequency characteristic F 4 of the parallel circuit of the inductor L 3 and the capacitor C 3 , which is symmetrically folded at the local oscillation frequency f LO as the center, is up-converted to the transmission lines 1 a and 1 b side by the local oscillation frequency f LO , and it is possible to make a two-band band-pass characteristic F 4 ′ with the center at the local oscillation frequency f LO appear on the transmission lines 1 a and 1 b side. [0085] Considering the frequency f BB2 of the frequency-converted signals B 2 + and B 2 − , both when the frequency f RF of the input signals T + and T − is apart from f LO ±f R and when the frequency f RF of the input signals T + and T − is close to f LO ±f R , the frequency f BB2 of the frequency-converted signals B 2 + and B 2 − becomes high, and the impedance Z of the parallel circuit of the inductor L 3 and the capacitor C 3 is decreased. [0086] For this reason, for the frequency f BB2 , a large change does not occur in the impedance Z on the frequency characteristic F 4 of the parallel circuit of the inductor L 3 and the capacitor C 3 , and for the frequency f BB1 , a large change occurs in the impedance Z on the frequency characteristic F 4 of the parallel circuit of the inductor L 3 and the capacitor C 3 . Therefore, when the total impedance characteristic of the output terminals of the passive mixer 12 is observed from the transmission lines 1 a and 1 b side, the change of the impedance Z can be ignored on the frequency characteristic F 4 in the frequency f BB2 , and only the change of the impedance Z on the frequency characteristic F 4 in the frequency f BB1 can be dominantly observed as the total change of the impedance Z. [0087] As a result, by using the parallel circuit of the inductor L 3 and the capacitor C 3 as the frequency response device 3 of the filter circuit shown in FIG. 7 , it is possible to obtain the band-pass characteristic F 4 ′ with f LO ±f R as the center with respect to transmission lines 1 a and 1 b. Because the local oscillation frequency f LO can be freely varied, the center frequency of the pass band can be also changed freely and precisely. Furthermore, because the Q factors of the inductor L 3 and the capacitor C 3 can be easily increased, a band-pass filter having an extremely narrow bandwidth can be constructed. In addition, because the passive mixer 12 can be easily formed by a semiconductor manufacturing process, it can be constructed on the same integrated circuit (IC) chip with the other circuits, which leads to a cost down without using an external filter device. Moreover, since it does not consume the electric power because it is a passive mixer, the filter function can be added without increasing the power consumption. As a result, by using the passive mixer 12 and the parallel circuit of the inductor L 3 and the capacitor C 3 , it is possible to realize a high-performance tunable band-pass filter with ease. [0088] FIG. 12 is a block diagram of a filter circuit according to an eighth embodiment of the present invention. [0089] As shown in FIG. 12 , the filter circuit according to the eighth embodiment includes a passive mixer 12 and frequency response devices 3 a and 3 b. The input terminals of the passive mixer 12 are connected to transmission lines 1 a and 1 b, respectively, and the output terminals of the passive mixer 12 are grounded via the frequency response devices 3 a and 3 b, respectively. [0090] The passive mixer 12 can generate frequency-converted signals B 3 + and B 3 − by mixing input signals T + and T − that are transmitted through the transmission lines 1 a and 1 b and local oscillation signals LO + and LO − , respectively. The passive mixer 12 can shift a frequency characteristic F of the frequency response devices 3 a and 3 b to high frequency by a local oscillation frequency f LO of the local oscillation signals LO + and LO − by outputting the frequency-converted signals B 3 + and B 3 − to the frequency response devices 3 a and 3 b, respectively, and apply the frequency characteristic F that is shifted to high frequency on the input signals T + and T − . [0091] In other words, by applying the frequency characteristic F of the frequency response devices 3 a and 3 b on the frequency-converted signals B 3 + and B 3 − , respectively, it is possible to apply a frequency characteristic F′ which is the frequency characteristic F shifted to high frequency by the local oscillation frequency f LO , on the input signals T + and T − . [0092] Therefore, for the input signals T + and T − , the passive mixer 12 and the frequency response devices 3 a and 3 b can operate as a filter circuit that has the frequency characteristic F′ which is the frequency characteristic F shifted to high frequency by the local oscillation frequency f LO . As a result, even when a high-frequency differential signal is input as the input signals T + and T − , the center frequency of the filter circuit can be precisely matched to a target value, and at the same time, it is possible to make the attenuation characteristic steep and increase the attenuation amount for the input signals T + and T − , and to increase the signal transmission speed while decreasing the common-mode noise. [0093] FIG. 13 is a block diagram of a filter circuit according to a ninth embodiment of the present invention. [0094] As shown in FIG. 13 , the filter circuit according to the ninth embodiment includes passive mixers 2 a and 2 b and frequency response devices 3 a and 3 b. The input terminals of the passive mixers 2 a and 2 b are connected to a transmission line 1 , and the output terminals of the passive mixers 2 a and 2 b are grounded via the frequency response devices 3 a and 3 b, respectively. The passive mixer 2 a can be used for quadrature phase signals, and the passive mixer 2 b can be used for an inphase signal. [0095] The passive mixers 2 a and 2 b can generate frequency-converted signals B 4 + and B 4 − by mixing an input signal T that is transmitted through the transmission line 1 and local oscillation signals LO_Q and LO_I, respectively. As for the local oscillation signals LO_Q and LO_I, signals having the same frequency with phases shifted by 90 degrees can be used. [0096] The passive mixers 2 a and 2 b can shift the frequency characteristic F of the frequency response devices 3 a and 3 b to high frequency by a local oscillation frequency f LO of the local oscillation signals LO_Q and LO_I by outputting the frequency-converted signals B 4 + and B 4 − to the frequency response devices 3 a and 3 b, respectively, and thus applying the frequency characteristic F that is shifted to high frequency on the input signal T. In other words, by applying the frequency characteristic F of the frequency response devices 3 a and 3 b on the frequency-converted signals B 4 + and B 4 − , respectively, it is possible to apply a frequency characteristic F′ which is the frequency characteristic F shifted to high frequency by the local oscillation frequency f LO , on the input signal T. [0097] Therefore, for the quadrature phase components and the inphase components of the input signal T, the passive mixers 2 a and 2 b and the frequency response devices 3 a and 3 b can operate as a filter circuit that has the frequency characteristic F′ which is the frequency characteristic F shifted to high frequency by the local oscillation frequency f LO . As a result, even when the frequency of the input signal T is high, the quadrature phase signals and the inphase signals used in a wireless communication apparatus can be handled, the center frequency of the filter circuit can be precisely matched to a target value, and at the same time, it is possible to make the attenuation characteristic steep and increase the attenuation amount for the input signal T. [0098] In a wireless communication circuit, the local oscillation signal LO is less often used in a single signal, and two systems of local oscillation signals LO_Q and LO_I with the phases shifted by 90 degrees are used in most cases. Although it is possible to construct the filter circuit shown in FIG. 1 , FIG. 6 , FIG. 7 , or FIG. 12 by extracting one side of the two systems, if only the one side of the two systems is extracted, the circuit structure becomes unbalanced, resulting in a failure in keeping 90 degrees of the phase difference between the local oscillation signals LO_Q and LO_I or an occurrence of a difference in the voltage amplitudes. Therefore, as shown in FIG. 13 , an adverse effect on the other circuits can be prevented by taking the configuration of using both of the two systems of the local oscillation signals LO_Q and LO_I. [0099] FIG. 14 is a block diagram of a filter circuit according to a tenth embodiment of the present invention. [0100] As shown in FIG. 14 , the filter circuit according to the tenth embodiment includes passive mixers 12 a and 12 b and frequency response devices 3 a and 3 b. As for the passive mixers 12 a and 12 b, a differential passive mixer can be used. The passive mixer 12 a can be used for quadrature phase signals, and the passive mixer 12 b can be used for an inphase signal. The input terminals of the passive mixer 12 a are connected to transmission lines 1 a and 1 b, respectively, and the output terminals of the passive mixer 12 a are connected to each other via the frequency response device 3 a. Similarly, the input terminals of the passive mixer 12 b are connected to transmission lines 1 a and 1 b, respectively, and the output terminals of the passive mixer 12 b are connected to each other via the frequency response device 3 b. [0101] The passive mixer 12 a can generate frequency-converted signals B 5 Q+ and B 5 Q− by mixing input signals T + and T − that are transmitted through the transmission lines 1 a and 1 b, respectively, and local oscillation signals LO_Q + and LO_Q − , respectively. The passive mixer 12 b can generate frequency-converted signals B 5 Q+ and B 5 Q− by mixing input signals T + and T − that are transmitted through the transmission lines 1 a and 1 b, respectively, and local oscillation signals LO_I + and LO_I − , respectively. As for the local oscillation signals LO_Q + , LO_Q − , LO_I + , and LO_I − , signals having the same frequency with phases sequentially shifted by 90 degrees can be used. [0102] The passive mixer 12 a can shift a frequency characteristic F of the frequency response device 3 a to high frequency by a local oscillation frequency f LO of the local oscillation signals LO_Q + and LO_Q − by outputting the frequency-converted signals B 5 Q+ and B 5 Q− to both terminals of the frequency response device 3 a, respectively, and apply the frequency characteristic F that is shifted to high frequency on the input signals T + and T − . The passive mixer 12 b can shift a frequency characteristic F of the frequency response device 3 b to high frequency by a local oscillation frequency f LO of the local oscillation signals LO_I + and LO_I − by outputting the frequency-converted signals B 5 I+ and B 5 I− to both terminals of the frequency response device 3 b, respectively, and apply the frequency characteristic F that is shifted to high frequency on the input signals T + and T − . [0103] In other words, by applying the frequency characteristic F of the frequency response device 3 a on the frequency-converted signals B 5 Q+ and B 5 Q− , it is possible to apply a frequency characteristic F′ which is the frequency characteristic F shifted to high frequency by the local oscillation frequency f LO , on the quadrature phase components of the input signals T + and T − . Similarly, by applying the frequency characteristic F of the frequency response device 3 b on the frequency-converted signals B 5 I+ and B 5 I− , it is possible to apply a frequency characteristic F′ which is the frequency characteristic F shifted to high frequency by the local oscillation frequency f LO , on the inphase components of the input signals T + and T − . [0104] Therefore, for the quadrature phase components and the inphase components of the input signals T + and T − , the passive mixers 12 a and 12 b and the frequency response devices 3 a and 3 b can operate as a filter circuit that has the frequency characteristic F′ which is the frequency characteristic F shifted to high frequency by the local oscillation frequency f LO . As a result, even when high-frequency differential signals are input as the quadrature phase components and the inphase components of the input signals T + and T − , the center frequency of the filter circuit can be precisely matched to a target value, and at the same time, it is possible to make the attenuation characteristic steep and increase the attenuation amount for the input signals T + and T − , and to increase the signal transmission speed while decreasing the common-mode noise. [0105] FIG. 15 is a block diagram of a filter circuit according to an eleventh embodiment of the present invention. [0106] As shown in FIG. 15 , the filter circuit according to the eleventh embodiment includes passive mixers 12 a and 12 b and frequency response devices 13 a, 14 a, 13 b, and 14 b. The input terminals of the passive mixer 12 a are connected to transmission lines 1 a and 1 b, respectively, and the output terminals of the passive mixer 12 a are grounded via the frequency response devices 13 a and 14 a, respectively. Similarly, the input terminals of the passive mixer 12 b are connected to transmission lines 1 a and 1 b, respectively, and the output terminals of the passive mixer 12 b are grounded via the frequency response devices 13 b and 14 b, respectively. [0107] The passive mixer 12 a can generate frequency-converted signals B 6 Q+ and B 6 Q− by mixing input signals T + and T − that are transmitted through the transmission lines 1 a and 1 b, respectively, and local oscillation signals LO_Q + and LO_Q − , respectively. The passive mixer 12 a can shift a frequency characteristic F of the frequency response devices 13 a and 14 a to high frequency by a local oscillation frequency f LO of the local oscillation signals LO_Q + and LO_Q − by outputting the frequency-converted signals B 6 Q+ and B 6 Q− to the frequency response devices 13 a and 14 a, respectively, and apply the frequency characteristic F that is shifted to high frequency on the input signals T + and T − . [0108] The passive mixer 12 b can generate frequency-converted signals B 6 I+ and B 6 I− by mixing input signals T + and T − that are transmitted through the transmission lines 1 a and 1 b, respectively, and local oscillation signals LO_I + and LO_I − , respectively. The passive mixer 12 b can shift a frequency characteristic F of the frequency response devices 13 b and 14 b to high frequency by a local oscillation frequency f LO of the local oscillation signals LO_I + and LO_I − by outputting the frequency-converted signals B 6 I+ and B 6 I− to the frequency response devices 13 b and 14 b, respectively, and apply the frequency characteristic F that is shifted to high frequency on the input signals T + and T − . [0109] In other words, by applying the frequency characteristic F of the frequency response devices 13 a and 14 a on the frequency-converted signals B 6 Q+ and B 6 Q− , it is possible to apply a frequency characteristic F′ which is the frequency characteristic F shifted to high frequency by the local oscillation frequency f LO , on the quadrature phase components of the input signals T + and T − . Similarly, by applying the frequency characteristic F of the frequency response devices 13 b and 14 b on the frequency-converted signals B 6 I+ and B 6 I− , it is possible to apply a frequency characteristic F′ which is the frequency characteristic F shifted to high frequency by the local oscillation frequency f LO , on the inphase components of the input signals T + and T − . [0110] Therefore, for the quadrature phase components and the inphase components of the input signals T + and T − , the passive mixers 12 a and 12 b and the frequency response devices 13 a, 14 a, 13 b, and 14 b can operate as a filter circuit that has the frequency characteristic F′ which is the frequency characteristic F shifted to high frequency by the local oscillation frequency f LO . As a result, even when high-frequency differential signals are input as the quadrature phase components and the inphase components of the input signals T + and T − , the center frequency of the filter circuit can be precisely matched to a target value, and at the same time, it is possible to make the attenuation characteristic steep and increase the attenuation amount for the input signal T + and T − , and to increase the signal transmission speed while decreasing the common-mode noise. [0111] FIG. 16 is a block diagram of a filter circuit according to a twelfth embodiment of the present invention. [0112] As shown in FIG. 16 , the filter circuit according to the twelfth embodiment includes passive mixers 22 a and 22 b and a frequency response device 3 . As for the passive mixers 22 a and 22 b, a single-phase passive mixer can be used. The passive mixer 22 a can be used for quadrature phase signals, and the passive mixer 22 b can be used for an inphase signal. The input terminal of the passive mixer 22 a is connected to a transmission line 1 b, and the input terminal of the passive mixer 22 b is connected to a transmission line 1 a. The output terminals of the passive mixers 22 a and 22 b are connected to each other via the frequency response device 3 . [0113] The passive mixer 22 a can generate a frequency-converted signal B 7 + by mixing an input signal T − that is transmitted through the transmission line 1 b and local oscillation signals LO_Q + and LO_Q − , respectively. Similarly, the passive mixer 22 b can generate a frequency-converted signal B 7 − by mixing an input signal T + that is transmitted through the transmission line 1 a and local oscillation signals LO_I + and LO_I − . [0114] The passive mixers 22 a and 22 b can shift a frequency characteristic F of the frequency response device 3 to high frequency by a local oscillation frequency f LO of the local oscillation signals LO_Q + , LO_Q − , LO_I + , and LO_I − by outputting the frequency-converted signals B 7 + and B 7 − to both terminals of the frequency response device 3 , respectively, and apply the frequency characteristic F that is shifted to high frequency on the input signals T + and T − . [0115] In other words, by applying the frequency characteristic F of the frequency response device 3 on the frequency-converted signals B 7 + and B 7 − , it is possible to apply a frequency characteristic F′ which is the frequency characteristic F shifted to high frequency by the local oscillation frequency f LO , on the quadrature phase components of the input signals T + and T − . [0116] Therefore, for the quadrature phase components and the inphase components of the input signals T + and T − , the passive mixers 22 a and 22 b and the frequency response device 3 can operate as a filter circuit that has the frequency characteristic F′ which is the frequency characteristic F shifted to high frequency by the local oscillation frequency f LO , while preventing a part of the frequency characteristic from being canceled between the passive mixers 22 a and 22 b. As a result, even when high-frequency differential signals are input as the quadrature phase components and the inphase components of the input signals T + and T − , the center frequency of the filter circuit can be precisely matched to a target value, and at the same time, it is possible to make the attenuation characteristic steep and increase the attenuation amount for the input signals T + and T − , and to increase the signal transmission speed while decreasing the common-mode noise. [0117] In the configuration shown in FIG. 14 , the passive mixers 12 a and 12 b are connected to both the transmission lines 1 a and 1 b, respectively. Therefore, with the configuration shown in FIG. 14 , even though it is a part of it, the frequency characteristic of the passive mixer 12 a and the frequency characteristic of the passive mixer 12 b are canceled with each other, so that the frequency characteristic becomes slightly mild, resulting in a degradation of the steepness of the attenuation characteristic compared to the configuration shown in FIG. 7 . On the other hand, in the configuration shown in FIG. 16 , the passive mixers 22 a and 22 b are connected to one side of the transmission lines 1 a and 1 b, respectively. For this reason, with the configuration shown in FIG. 16 , the frequency characteristics of the passive mixers 22 a and 22 b are not canceled with each other, so that it is possible to prevent the frequency characteristic from being mild, making it possible to realize a steep attenuation characteristic similar to the case in the configuration shown in FIG. 7 . [0118] FIG. 17 is a block diagram of a filter circuit according to a thirteenth embodiment of the present invention. [0119] As shown in FIG. 17 , the filter circuit according to the thirteenth embodiment includes passive mixers 22 a and 22 b and frequency response devices 3 a and 3 b. The input terminal of the passive mixer 22 a is connected to a transmission line 1 b, and the input terminal of the passive mixer 22 b is connected to a transmission line 1 a. The output terminals of the passive mixers 22 a and 22 b are grounded via the frequency response devices 3 a and 3 b, respectively. [0120] The passive mixer 22 a can generate a frequency-converted signal B 8 + by mixing an input signal T that is transmitted through the transmission line 1 b and local oscillation signals LO_Q + and LO_Q − , respectively. Similarly, the passive mixer 22 b can generate a frequency-converted signal B 8 − by mixing an input signal T + that is transmitted through the transmission line 1 a and local oscillation signals LO_I + and LO_I − . [0121] The passive mixers 22 a and 22 b can shift a frequency characteristic F of the frequency response device 3 a and 3 b to high frequency by a local oscillation frequency f LO of the local oscillation signals LO_Q + , LO_Q − , LO_I + , and LO_I − by outputting the frequency-converted signals B 8 + and B 8 − to the frequency response devices 3 a and 3 b, respectively, and apply the frequency characteristic F that is shifted to high frequency on the input signals T + and T − . [0122] In other words, by applying the frequency characteristic F of the frequency response devices 3 a and 3 b on the frequency-converted signals B 8 + and B 8 − , it is possible to apply a frequency characteristic F′ which is the frequency characteristic F shifted to high frequency by the local oscillation frequency f LO , on the quadrature phase components of the input signals T + and T − . [0123] Therefore, for the quadrature phase components and the inphase components of the input signals T + and T − , the passive mixers 22 a and 22 b and the frequency response devices 3 a and 3 b can operate as a filter circuit that has the frequency characteristic F′ which is the frequency characteristic F shifted to high frequency by the local oscillation frequency f LO , while preventing a part of the frequency characteristic from being canceled between the passive mixers 22 a and 22 b. As a result, even when high-frequency differential signals are input as the quadrature phase components and the inphase components of the input signals T + and T − , the center frequency of the filter circuit can be precisely matched to a target value, and at the same time, it is possible to make the attenuation characteristic steep and increase the attenuation amount for the input signals T + and T − , and to increase the signal transmission speed while decreasing the common-mode noise. [0124] In the configuration shown in FIG. 15 , the passive mixers 12 a and 12 b are connected to both the transmission lines 1 a and 1 b, respectively. Therefore, with the configuration shown in FIG. 15 , even though it is a part of it, the frequency characteristic of the passive mixer 12 a and the frequency characteristic of the passive mixer 12 b are canceled with each other, so that the frequency characteristic becomes slightly mild, resulting in a degradation of the steepness of the attenuation characteristic compared to the configuration shown in FIG. 12 . On the other hand, in the configuration shown in FIG. 17 , the passive mixers 22 a and 22 b are connected to one side of the transmission lines 1 a and 1 b, respectively. For this reason, with the configuration shown in FIG. 17 , the frequency characteristics of the passive mixers 22 a and 22 b are not canceled with each other, so that it is possible to prevent the frequency characteristic from being mild, making it possible to realize a steep attenuation characteristic similar to the case in the configuration shown in FIG. 12 . [0125] FIG. 18 is a block diagram of a passive mixer applied to a filter circuit according to a fourteenth embodiment of the present invention. [0126] In FIG. 18 , the passive mixer 2 shown in FIG. 1 is configured with a switch SW 1 . As for the switch SW 1 , a mechanical switch, such as a microelectoromechanical system (MEMS) switch and a relay switch, or a semiconductor switch, such as a field effect transistor, can be used. [0127] A first terminal of the switch SW 1 is connected to the transmission line 1 , and a second terminal of the switch SW 1 is connected to the frequency response device 3 . The input signal T and the local oscillation signal LO are mixed by switching on and off the switch SW 1 with the local oscillation signal LO, so that the mixed signal is output to the frequency response device 3 . [0128] The passive mixers 2 a and 2 b shown in FIG. 13 can be respectively configured with the switch SW 1 . [0129] FIG. 19 is a block diagram of a passive mixer applied to a filter circuit according to a fifteenth embodiment of the present invention. [0130] In FIG. 19 , the passive mixer 2 ′ shown in FIG. 6 is configured with a switch SW 2 . A first terminal of the switch SW 2 is connected to the transmission line 1 , one of second terminals of the switch SW 2 is connected to the frequency response device 3 a, and other of the second terminals of the switch SW 2 is connected to the frequency response device 3 b. The input signal T and the local oscillation signal LO are mixed by switching on and off the switch SW 2 with the local oscillation signal LO, so that the mixed signal is output to the frequency response devices 3 a and 3 b. [0131] FIG. 20 is a block diagram of a passive mixer applied to a filter circuit according to a sixteenth embodiment of the present invention. [0132] In FIG. 20 , the passive mixer 12 shown in FIG. 7 is configured with switches SW 11 and SW 12 . A first terminal of the switch SW 11 is connected to the transmission line 1 b, and a second terminal of the switch SW 11 is connected to a first terminal of the frequency response device 3 . Similarly, a first terminal of the switch SW 12 is connected to the transmission line 1 a, and a second terminal of the switch SW 12 is connected to a second terminal of the frequency response device 3 . The input signals T + and T − and the local oscillation signals LO + and LO − are mixed by switching on and off the switches SW 11 and SW 12 with the local oscillation signals LO + and LO − , respectively, so that the mixed signal is output to the frequency response device 3 . [0133] Alternatively, the passive mixer 12 shown in FIG. 12 can be configured with the switches SW 11 and SW 12 . In this case, the first terminal of the switch SW 11 can be connected to the transmission line 1 b, and the second terminal of the switch SW 11 can be connected to the frequency response device 3 a. Similarly, the first terminal of the switch SW 12 can be connected to the transmission line 1 a, and the second terminal of the switch SW 12 can be connected to the frequency response device 3 b. Then, the input signals T + and T − and the local oscillation signals LO + and LO − are mixed by switching on and off the switches SW 12 and SW 11 with the local oscillation signals LO + and LO − , respectively, so that the mixed signals are output to the frequency response devices 3 a and 3 b, respectively. [0134] Furthermore, the passive mixers 12 a and 12 b shown in FIGS. 14 and 15 can be respectively configured with the passive mixer 12 shown in FIG. 20 . In this case, the local oscillation signals LO_Q + and LO_Q can be used for a first passive mixer 12 instead of the local oscillation signals LO + and LO − , and the local oscillation signals LO_I + and LO_I − can be used for a second passive mixer 12 instead of the local oscillation signals LO + and LO − . [0135] Moreover, the passive mixers 22 a and 22 b shown in FIGS. 16 and 17 can be configured with a common passive mixer 12 shown in FIG. 20 . In this case, the local oscillation signals LO_I + and LO_I − or the local oscillation signals LO_Q + and LO_Q can be used for the passive mixer 12 instead of the local oscillation signals LO + and LO − . [0136] FIG. 21 is a block diagram of a passive mixer applied to a filter circuit according to a seventeenth embodiment of the present invention. [0137] In FIG. 21 , the passive mixer 12 shown in FIG. 7 is configured with switches SW 21 and SW 22 . A first terminal of the switch SW 21 is connected to the transmission line 1 b, one of second terminals of the switch SW 21 is connected to a first terminal of the frequency response device 3 , and other of the second terminals of the switch SW 21 is connected to a second terminal of the frequency response device 3 . A first terminal of the switch SW 22 is connected to the transmission line 1 a, one of second terminals of the switch SW 22 is connected to the first terminal of the frequency response device 3 , and other of the second terminals of the switch SW 22 is connected to the second terminal of the frequency response device 3 . The input signals T + and T − and the local oscillation signals LO + and LO − are mixed by switching on and off the switches SW 22 and SW 21 with the local oscillation signals LO + and LO − , respectively, so that the mixed signal is output to the frequency response device 3 . [0138] Alternatively, the passive mixer 12 shown in FIG. 12 can be configured with the switches SW 21 and SW 22 . In this case, the first terminal of the switch SW 21 can be connected to the transmission line 1 b, one of the second terminals of the switch SW 21 can be connected to a first terminal of the frequency response device 3 a, and other of the second terminals of the switch SW 21 can be connected to a first terminal of the frequency response device 3 b. Similarly, the first terminal of the switch SW 22 can be connected to the transmission line 1 a, one of the second terminals of the switch SW 22 can be connected to the first terminal of the frequency response device 3 a, and other of the second terminals of the switch SW 22 can be connected to the first terminal of the frequency response device 3 b. Then, the input signals T + and T and the local oscillation signals LO + and LO − are mixed by switching on and off the switches SW 22 and SW 21 with the local oscillation signals LO + and LO − , respectively, so that the mixed signals are output to the frequency response devices 3 a and 3 b, respectively. [0139] Furthermore, the passive mixers 12 a and 12 b shown in FIGS. 14 and 15 can be respectively configured with the passive mixer 12 shown in FIG. 21 . In this case, the local oscillation signals LO_Q + and LO_Q − can be used for a first passive mixer 12 instead of the local oscillation signals LO + and LO − , and the local oscillation signals LO_I + and LO_I − can be used for a second passive mixer 12 instead of the local oscillation signals LO + and LO − . [0140] Moreover, the passive mixers 22 a and 22 b shown in FIGS. 16 and 17 can be configured with a common passive mixer 12 shown in FIG. 21 . In this case, the local oscillation signals LO_I + and LO_I − or the local oscillation signals LO_Q + and LO_Q − can be used for the passive mixer 12 instead of the local oscillation signals LO + and LO − . [0141] FIG. 22 is a block diagram of a passive mixer applied to a filter circuit according to an eighteenth embodiment of the present invention. [0142] In FIG. 22 , the passive mixer 2 shown in FIG. 1 is configured with a field effect transistor M 1 . The drain of the FET M 1 is connected to the transmission line 1 , and the source of the field effect transistor M 1 is connected to the frequency response device 3 . The local oscillation signal LO is input to the gate of the field effect transistor M 1 , and then, the input signal T and the local oscillation signal LO are mixed by switching on and off the field effect transistor M 1 with the local oscillation signal LO. [0143] Furthermore, the passive mixers 2 a and 2 b shown in FIG. 13 can be respectively configured with the field effect transistor M 1 . [0144] FIG. 23 is a block diagram of a passive mixer applied to a filter circuit according to a nineteenth embodiment of the present invention. [0145] In FIG. 23 , the passive mixer 2 ′ shown in FIG. 6 is configured with field effect transistors M 2 and M 3 . The drains of the field effect transistors M 2 and M 3 are commonly connected to the transmission line 1 , the source of the field effect transistor M 2 is connected to the frequency response device 3 a, and the source of the field effect transistor M 3 is connected to the frequency response device 3 b. The input signal T and the local oscillation signals LO + and LO − are mixed inputting the local oscillation signals LO + and LO − to the gates of the field effect transistors M 2 and M 3 , respectively, thus switching on and off the field effect transistors M 2 and M 3 with the local oscillation signals LO + and LO − , respectively, so that the mixed signals are output to the frequency response devices 3 a and 3 b, respectively. [0146] FIG. 24 is a block diagram of a passive mixer applied to a filter circuit according to a twentieth embodiment of the present invention. [0147] In FIG. 24 , the passive mixer 12 shown in FIG. 7 is configured with field effect transistors M 11 and M 12 . The drain of the field effect transistor M 11 is connected to the transmission line 1 b, and the source of the field effect transistor M 11 is connected to a first terminal of the frequency response device 3 . Similarly, the drain of the field effect transistor M 12 is connected to the transmission line 1 a, and the source of the field effect transistor M 12 is connected to a second terminal of the frequency response device 3 . The input signals T + and T − and the local oscillation signals LO + and LO − are mixed by inputting the local oscillation signals LO + and LO − to the gates of the field effect transistors M 11 and M 12 , respectively, thus switching on and off the field effect transistors M 11 and M 12 with the local oscillation signals LO + and LO − , respectively, so that the mixed signal is output to the frequency response device 3 . [0148] Alternatively, the passive mixer 12 shown in FIG. 12 can be configured with the field effect transistors M 11 and M 12 . In this case, the drain of the field effect transistor M 11 can be connected to the transmission line 1 b, and the source of the field effect transistor M 11 can be connected to a first terminal of the frequency response device 3 a. Similarly, the drain of the field effect transistor M 12 can be connected to the transmission line 1 a, and the source of the field effect transistor M 12 can be connected to a first terminal of the frequency response device 3 b. [0149] Furthermore, the passive mixers 12 a and 12 b shown in FIGS. 14 and 15 can be respectively configured with the passive mixer 12 shown in FIG. 24 . Moreover, the passive mixers 22 a and 22 b shown in FIGS. 16 and 17 can be configured with a common passive mixer 12 shown in FIG. 24 . [0150] FIG. 25 is a block diagram of a passive mixer applied to a filter circuit according to a twenty-first embodiment of the present invention. [0151] In FIG. 25 , the passive mixer 12 shown in FIG. 7 is configured with field effect transistors M 21 to M 24 . The drains of the field effect transistors M 21 and M 22 are commonly connected to the transmission line 1 b, and the drains of the field effect transistors M 23 and M 24 are commonly connected to the transmission line 1 a. The sources of the field effect transistors M 21 and M 23 are commonly connected to a first terminal of the frequency response device 3 , and the sources of the field effect transistors M 22 and M 24 are commonly connected to a second terminal of the frequency response device 3 . The gates of the field effect transistors M 21 and M 24 are connected to each other, and the gates of the field effect transistors M 22 and M 23 are connected to each other. [0152] The input signals T + and T − and the local oscillation signals LO + and LO − are mixed by commonly inputting the local oscillation signal LO + to the gates of the field effect transistors M 22 and M 23 and commonly inputting the local oscillation signal LO − to the gates of the field effect transistors M 21 and M 24 , thus switching on and off the field effect transistors M 21 to M 24 with the local oscillation signals LO + and LO − , so that the mixed signal is output to the frequency response device 3 . [0153] Alternatively, the passive mixer 12 shown in FIG. 12 can be configured with the field effect transistors M 21 to M 24 . Furthermore, the passive mixers 12 a and 12 b shown in FIGS. 14 and 15 can be respectively configured with the passive mixer 12 shown in FIG. 25 . [0154] FIG. 26 is a block diagram of a passive mixer applied to a filter circuit according to a twenty-second embodiment of the present invention. [0155] In FIG. 26 , the passive mixer 22 a shown in FIG. 16 is configured with field effect transistors M 31 and M 32 , and the passive mixer 22 b is configured with field effect transistors M 33 and M 34 . The drains of the field effect transistors M 31 and M 32 are commonly connected to the transmission line 1 b, and the drains of the field effect transistors M 33 and M 34 are commonly connected to the transmission line 1 a. The sources of the field effect transistors M 31 and M 33 are commonly connected to a first terminal of the frequency response device 3 , and the sources of the field effect transistors M 32 and M 34 are commonly connected to a second terminal of the frequency response device 3 . [0156] The input signals T + and T − and the local oscillation signals LO_Q + , LO_Q − , LO_I + , and LO_I − are mixed by commonly inputting the local oscillation signals LO_Q + and LO_Q − to the gates of the field effect transistors M 32 and M 31 , respectively, and commonly inputting the local oscillation signals LO_I + and LO_I − to the gates of the field effect transistors M 33 and M 34 , respectively, thus switching on and off the field effect transistors M 31 to M 34 with the local oscillation signals LO_Q + , LO_Q − , LO_I + , and LO_I − , so that the mixed signal is output to the frequency response device 3 . [0157] Alternatively, the passive mixers 22 a and 22 b shown in FIG. 17 can be configured with the passive mixers 22 a and 22 b shown in FIG. 26 . Furthermore, the passive mixers 22 a and 22 b shown in FIGS. 16 and 17 can be configured with a common passive mixer 12 shown in FIG. 26 . [0158] FIG. 27 is a block diagram of a wireless transmitter according to a twenty-third embodiment of the present invention to which the filter circuit according to the present invention is applied. [0159] As shown in FIG. 27 , the wireless transmitter according to the twenty-third embodiment includes up-converters 103 and 104 for performing an up-convert of a transmission signal. The input terminals of the up-converters 103 and 104 are connected to filter devices 101 and 102 , respectively, and the output terminals of the up-converters 103 and 104 are commonly connected to a transmission antenna 108 via a filter circuit 105 , a variable gain amplifier 106 , and a power amplifier 107 in a sequential manner. As for the filter circuit 105 , the filter circuit shown in any one of FIGS. 1 , 6 , 7 , and 12 to 26 can be used. [0160] When the filter circuit shown in any one of FIGS. 1 , 6 , 13 , 18 , 19 , 22 , and 23 is used as the filter circuit 105 , the input terminal and the output terminal of the filter circuit 105 are connected to the transmission line 1 shown in the figures. Furthermore, when the filter circuit shown in any one of FIGS. 7 , 12 , 14 , 15 , 16 , 17 , 20 , 21 , 24 , 25 , and 26 is used as the filter circuit 105 , the input terminal and the output terminal of the filter circuit 105 are connected to the transmission lines 1 a and 1 b shown in the figures. [0161] An inphase component I of a baseband signal is input to the up-converter 103 after an unnecessary frequency component is eliminated at the filter device 101 . The inphase component I of the baseband signal is up-converted by being mixed with a local oscillation signal L 1 at the up-converter 103 , so that the inphase component I of a radio-frequency signal is generated. The quadrature phase component Q of the baseband signal is input to the up-converter 104 after an unnecessary frequency component is eliminated at the filter device 102 . The quadrature phase component Q of the baseband signal is up-converted by being mixed with a local oscillation signal LQ at the up-converter 104 , so that the quadrature phase component Q of a radio-frequency signal is generated. As for the local oscillation signals L 1 and LQ, signals having the same frequency with phases shifted by 90 degrees can be used. [0162] The inphase component I and the quadrature phase component Q of the radio-frequency signal respectively generated at the up-converters 103 and 104 are amplified at the variable gain amplifier 106 and the power amplifier 107 after an unnecessary frequency component is eliminated at the filter circuit 105 , respectively, and then transmitted through the space via the transmission antenna 108 . [0163] The up-converters 103 and 104 are used for up-converting the transmission signal, however, the passive mixer included in the filter circuit 105 is used for shifting the frequency characteristic of the frequency response device included in the filter circuit 105 to high frequency without up-converting the transmission signal. [0164] Conventionally, an LC filter or a surface acoustic wave (SAW) filter is used in the filter circuit 105 . However, in the LC filter, it is hard to precisely match the resonant frequency to a target value by an electronic control, to make the attenuation characteristic steep, and to increase the attenuation amount. In the SAW filter, it is not possible to vary the center frequency and to mount it on the same IC chip together with the other circuits. On the other hand, if the filter circuit shown in any one of FIGS. 1 , 6 , 7 , and 12 to 26 is used as the filter circuit 105 , it is possible to construct a filter circuit having a steep attenuation characteristic and a large attenuation amount on the same IC chip with the other circuits, so that an enhancement of performance can be realized at low cost. [0165] Although a configuration in which the filter circuit 105 is arranged on the input side of the variable gain amplifier 106 in the twenty-third embodiment shown in FIG. 27 , the filter circuit 105 can be arranged on the output side of the variable gain amplifier 106 or on the output side of the power amplifier 107 . Furthermore, the filter circuit 105 can be arranged on the transmission line of a radio-frequency band, on the transmission line of an intermediate-frequency band, or on the transmission line of a baseband. In addition, the transmission line of the transmission signal can be a single-phase transmission line or a differential transmission line. [0166] FIG. 28 is a block diagram of a wireless receiver according to a twenty-fourth embodiment of the present invention to which the filter circuit according to the present invention is applied. [0167] As shown in FIG. 28 , the wireless receiver according to the twenty-fourth embodiment includes down-converters 204 and 205 for performing a down-convert of a reception signal. The input terminals of the down-converters 204 and 205 are commonly connected to a reception antenna 201 via a filter circuit 203 and a low-noise amplifier 202 . The output terminals of the down-converters 204 and 205 are connected to variable gain amplifiers 208 and 209 via filter devices 206 and 207 , respectively. As for the filter circuit 203 , the filter circuit shown in any one of FIGS. 1 , 6 , 7 , and 12 to 26 can be used. [0168] When the filter circuit shown in any one of FIGS. 1 , 6 , 13 , 18 , 19 , 22 , and 23 is used as the filter circuit 203 , the input terminal and the output terminal of the filter circuit 203 are connected to the transmission line 1 shown in the figures. Furthermore, when the filter circuit shown in any one of FIGS. 7 , 12 , 14 , 15 , 16 , 17 , 20 , 21 , 24 , 25 , and 26 is used as the filter circuit 203 , the input terminal and the output terminal of the filter circuit 203 are connected to the transmission lines 1 a and 1 b shown in the figures. [0169] A desired frequency component is selected by the filter circuit 203 from a radio-frequency signal received by the reception antenna 201 after being amplified at the low-noise amplifier 202 , and input to the down-converters 204 and 205 . The radio-frequency signal passed the filter circuit 203 is down-converted by being mixed with a local oscillation signal L 1 at the down-converter 204 , so that an inphase component I of a baseband signal is generated. Similarly, the radio-frequency signal passed the filter circuit 203 is down-converted at the down-converter 205 by being mixed with a local oscillation signal LQ, so that an quadrature phase component Q of the baseband signal is generated. [0170] The inphase component I and the quadrature phase component Q of the baseband signal respectively generated at the down-converters 204 and 205 are amplified at the variable gain amplifiers 208 and 209 after an unnecessary frequency component is eliminated at the filter devices 206 and 207 , respectively. [0171] The down-converters 204 and 205 are used for down-converting the reception signal, however, the passive mixer included in the filter circuit 203 is used for shifting the frequency characteristic of the frequency response device included in the filter circuit 203 to high frequency without down-converting the reception signal. [0172] Conventionally, an LC filter or a SAW filter is used in the filter circuit 203 . However, in the LC filter, it is hard to precisely match the resonant frequency to a target value by an electronic control, to make the attenuation characteristic steep, and to increase the attenuation amount. In the SAW filter, it is not possible to vary the center frequency and to mount it on the same IC chip together with the other circuits. On the other hand, if the filter circuit shown in any one of FIGS. 1 , 6 , 7 , and 12 to 26 is used as the filter circuit 203 , it is possible to construct a filter circuit having a steep attenuation characteristic and a large attenuation amount on the same IC chip with the other circuits, so that an enhancement of performance can be realized at low cost. [0173] Although a configuration in which the filter circuit 203 is arranged on the output side of the low-noise amplifier 202 in the twenty-fourth embodiment shown in FIG. 28 , the filter circuit 203 can be arranged on the input side of the low-noise amplifier 202 . Furthermore, the filter circuit 203 can be arranged on the transmission line of a radio-frequency band, on the transmission line of an intermediate-frequency band, or on the transmission line of a baseband. In addition, the transmission line of the reception signal can be a single-phase transmission line or a differential transmission line. [0174] Additional advantages and modifications will readily occur to those skilled in the art. Therefore, the invention in its broader aspects is not limited to the specific details and representative embodiments shown and described herein. Accordingly, various modifications may be made without departing from the spirit or scope of the general inventive concept as defined by the appended claims and their equivalents.

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